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LTC3565 1.25A, 4MHz, Synchronous Step-Down DC/DC Converter FEATURES

DESCRIPTION
The LTC(R)3565 is a constant frequency, synchronous step-down DC/DC converter. Intended for medium power applications, it operates from a 2.5V to 5.5V input voltage range and has a user configurable operating frequency up to 4MHz, allowing the use of tiny, low cost capacitors and inductors 1mm or less in height. The output voltage is adjustable from 0.6V to 5.5V. Internal synchronous power switches provide high efficiency. The LTC3565's current mode architecture and external compensation allow the transient response to be optimized over a wide range of loads and output capacitors. The LTC3565 can be configured for automatic power saving Burst Mode operation (IQ = 40A) to reduce gate charge losses when the load current drops below the level required for continuous operation. For reduced noise and RF interference, the SYNC/MODE pin can be configured to skip pulses or provide forced continuous operation. To further maximize battery life, the P-channel MOSFET is turned on continuously in dropout (100% duty cycle). In shutdown, the device draws <1A.
, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6580258, 6498466, 6611131.

High Efficiency: Up to 95% VIN Range: 2.5V to 5.5V High Frequency Operation: Up to 4MHz Selectable Low Ripple (Typical 25mVp-p) Burst Mode(R) Operation: IQ = 40A Stable with Ceramic Capacitors Uses Tiny Capacitors and Inductor Low RDS(ON) Internal Switches: 0.15 Current Mode Operation for Excellent Line and Load Transient Response Short-Circuit Protected Low Dropout Operation: 100% Duty Cycle Low Shutdown Current: IQ 1A Output Voltages from 0.6V to 5V Synchronizable to External Clock Supports Pre-Biased Outputs Small 10-Lead (3mm x 3mm) DFN or MSOP Package
APPLICATIONS

Notebook Computers Digital Cameras Cellular Phones Handheld Instruments Board Mounted Power Supplies
TYPICAL APPLICATION
Step-Down 2.5V/1.25A Regulator
VIN 2.5V TO 5.5V 22F EFFICIENCY (%) SYNC/MODE SVIN PVIN RUN PGOOD ITH RT 12.1k VFB GND 680pF 191k 22pF LTC3565 SW 2.2H VOUT 2.5V 1.25A 22F
Efficiency and Power Loss vs Output Current
100 90 80 70 60 50 40 30 20 VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 10 100 1000 OUTPUT CURRENT (mA) 0.01 0.1 POWER LOSS (W) 1
0.001
931k 294k
10 0 0.1
0.0001 10000
3565 TA01b
3565 TA01a
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LTC3565 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PVIN, SVIN Voltages ..................................... -0.3V to 6V VFB, ITH, RT Voltages ....................-0.3V to (VIN + 0.3V) SYNC/MODE, PGOOD Voltage .......-0.3V to (VIN + 0.3V) SW Voltage (DC) ...........................-0.3V to (VIN + 0.3V) RUN Voltage ............................................... -0.3V to 6V
Operating Junction Temperature Range (Notes 2, 5, 8) ........................................ -40C to 125C Storage Temperature Range................... -65C to 125C Lead Temperature (Soldering, 10 sec) .................. 300C
PIN CONFIGURATION
TOP VIEW RT RUN SYNC/MODE SW GND 1 2 3 4 5 11 10 ITH 9 VFB 8 PGOOD 7 SVIN 6 PVIN TOP VIEW RT RUN SYNC/MODE SW GND 1 2 3 4 5 10 9 8 7 6 ITH VFB PGOOD SVIN PVIN
11
DD PACKAGE 10-LEAD (3mm x 3mm) PLASTIC DFN TJMAX = 125C, JA = 43C/W, JC = 7.5C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
MSE PACKAGE 10-LEAD PLASTIC MSOP TJMAX = 125C, JA = 40C/W, JC = 10C/W EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3565EDD#PBF LTC3565IDD#PBF LTC3565EMSE#PBF LTC3565IMSE#PBF TAPE AND REEL LTC3565EDD#TRPBF LTC3565IDD#TRPBF LTC3565EMSE#TRPBF LTC3565IMSE#TRPBF PART MARKING* LTDNR LTDNR LTDVJ LTDVJ PACKAGE DESCRIPTION 10-Lead (3mm x 3mm) Plastic DFN 10-Lead (3mm x 3mm) Plastic DFN 10-Lead Plastic MSOP 10-Lead Plastic MSOP TEMPERATURE RANGE -40C to 125C -40C to 125C -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V, RT = 125k unless otherwise specified. (Note 2)
SYMBOL VIN IFB VFB VLINEREG VLOADREG gm(EA) PARAMETER Operating Voltage Range Feedback Pin Input Current Feedback Voltage Reference Voltage Line Regulation Output Voltage Load Regulation Error Amplifier Transconductance (Note 3) (Note 3) VIN = 2.5V to 5V ITH = 0.55V to 0.9V ITH Pin Load = 5A (Note 3)

ELECTRICAL CHARACTERISTICS
CONDITIONS
MIN 2.5 0.588
TYP
MAX 5.5 50
UNITS V nA V %/V % S
0.6 0.04 0.02 300
0.612 0.2 0.2
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LTC3565 ELECTRICAL CHARACTERISTICS
SYMBOL IS PARAMETER Input DC Supply Current (Note 4) Active Mode Sleep Mode Shutdown Oscillator Frequency Synchronization Frequency Peak Switch Current Limit Top Switch On-Resistance Bottom Switch On-Resistance ISW(LKG) VRUN IRUN VUVLO PGOOD RPGOOD PGOOD Blanking VSYNC-MODE tSOFT-START Pulse Skip Force Continuous Burst 10% to 90% of Regulation Switch Leakage Current RUN Threshold RUN Leakage Current Undervoltage Lockout Threshold Power Good Threshold Power Good Pull-Down On-Resistance VFB Step from 0V to 0.6V VFB Step from 0.6V to 0V 1.1 VIN - 0.75 0.6 VIN Ramping Down V FB Ramping Up from 0.45V to 0.6V VFB Ramping Down from 0.69V to 0.6V
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V, RT = 125k unless otherwise specified. (Note 2)
CONDITIONS VSYNC/MODE = 3.6V, VFB = 0.55V VSYNC/MODE = 3.6V, VFB = 0.8V VRUN = 0V RT = 125k (Note 7) (Note 7) VFB = 0.5V MSE Package DD Package (Note 6) MSE Package DD Package (Note 6) VIN = 5V, VRUN = 0V, VFB = 0V

MIN
TYP 330 40 0.1 1.5
MAX 450 60 1 1.7 4 4 2.5 0.2 0.18 1 1.5 1 2.2
UNITS A A A MHz MHz MHz A A V A V % % s s V V V ms
fOSC fSYNC ILIM RDS(ON)
1.3 0.4 1.5
2.1 0.15 0.15 0.13 0.13 0.01
0.3
0.8 0.01 1.9 -7 7 15 40 105
20
0.63 VIN - 1.05 0.9 1.2
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3565E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3565I is guaranteed over the full -40C to 125C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated at junction temperatures greater than 125C. Note 3: The LTC3565 is tested in a feedback loop which servos VFB to the midpoint for the error amplifier (VITH = 0.7V). Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency.
Note 5: TJ is calculated from the ambient TA and power dissipation PD according to the following formulas: LTC3565EDD: TJ = TA + (PD * 43C/W) LTC3565EMS: TJ = TA + (PD * 120C/W) Note 6: Switch on-resistance is guaranteed by correlation to wafer level measurements and assured by design characterization and correlation with statistical process controls. Note 7: 4MHz operation is guaranteed by design but not production tested and is subject to duty cycle limitations (see Applications Information). Note 8: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
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LTC3565 TYPICAL PERFORMANCE CHARACTERISTICS
otherwise noted. Efficiency vs Input Voltage
100 IOUT = 100mA 90 80 EFFICIENCY (%) 70 60 50 40 VOUT = 1.8V 30 2.5 3.0 4.5 4.0 3.5 INPUT VOLTAGE(V) 5.0 5.5
3565 G01
TA = 25C, VIN = 3.6V, fO = 1MHz, unless
Efficiency vs Output Current
100 100 90 80 EFFICIENCY (%) 70 60 50 40 30 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10000
3565 G02
Efficiency vs Output Current
IOUT = 10mA
90 80
IOUT = 1mA
EFFICIENCY (%)
IOUT = 1.25A
70 60 50 40 30 20 10 0 0.1 VOUT = 1.8V 1
IOUT = 0.1mA
20 10 0 0.1 VOUT = 1.5V 1
VIN = 2.7V VIN = 3.6V VIN = 4.2V 10000
3565 G03
100 1000 10 OUTPUT CURRENT (mA)
10 100 1000 OUTPUT CURRENT (mA)
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) EFFICIENCY (%) Burst Mode OPERATION 95 94 93
Efficiency vs Frequency
1.00 4.7H 0.75 2.2H 92 91 90 89 1H VOUT ERROR (%) 0.50
Load Regulation
70 60 50 40 30 20 10 0 0.1
PULSE SKIP
Burst Mode OPERATION 0.25 PULSE SKIP 0.00 FORCED CONTINUOUS
FORCED CONTINUOUS
-0.25 VOUT = 1.8V VOUT = 1.8V 5
3565 G05
VOUT = 1.8V 1 10 100 1000 OUTPUT CURRENT (mA) 10000
3565 G04
88 0 1 3 2 FREQUENCY (MHz) 4
-0.50 0 200
400 600 800 1000 1200 1400 OUTPUT CURRENT(mA)
3565 G06
Line Regulation
0.6 0.4 VOUT ERROR (%) 0.2 0.0 -0.2 -0.4 -0.6 2.5 615 610 REFERENCE VOLTAGE (mV) 605 600 595 590 VOUT = 1.8V ILOAD = 400mA 3.0 4.5 4.0 3.5 INPUT VOLTAGE(V) 5.0 5.5
3565 G07
Reference Voltage vs Temperature
6 4 2 0 -2 -4
Frequency Variation vs Temperature
585 -50 -25
FREQUENCY VARIATION (%) 50 25 75 0 TEMPERATURE (C) 100 125
-6 -50
-25
50 25 75 0 TEMPERATURE(C)
100
125
3565 G08
3565 G09
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LTC3565 TYPICAL PERFORMANCE CHARACTERISTICS
otherwise noted. Frequency Variation vs Input Voltage
6 4 FREQUENCY VARIATION (%) 2 RDS(ON) () 0 -2 -4 0.05 -6 -8 2.5 0.0 2.5 MAIN SWITCH SYNCHRONOUS SWITCH 3.0 4.5 4.0 3.5 INPUT VOLTAGE (V) 5.0 5.5
3565 G10
TA = 25C, VIN = 3.6V, fO = 1MHz, unless
RDS(ON) vs Input Voltage
0.25 0.30 0.25 0.20 0.15 0.10 0.05
RDS(ON) vs Temperature
0.20
0.15
0.10
RDS(ON) ()
MAIN SWITCH SYNCHRONOUS SWITCH -25 50 25 75 0 TEMPERATURE (C) 100 125
3.0
4.5 4.0 3.5 INPUT VOLTAGE (V)
5.0
5.5
3565 G11
0.0 -50
3565 G12
Dynamic Supply Current vs Input Voltage
100 DYNAMIC SUPPLY CURRENT (mA) DYNAMIC SUPPLY CURRENT (mA) 100 FORCED CONTINUOUS
Dynamic Supply Current vs Temperature
2500
Switch Leakage vs Input Voltage
10
10
FORCED CONTINUOUS SWITCH LEAKAGE (pA)
2000 MAIN SWITCH 1500
1 PULSE SKIP 0.1 Burst Mode OPERATION
1 PULSE SKIP 0.1 Burst Mode OPERATION
1000 SYNCHRONOUS SWITCH
0.01 VOUT = 1.8V ILOAD = 0A 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 5.5
3565 G13
0.01 VOUT = 1.8V ILOAD = 0A
500
0.001 2.5
0.001 -50
0 -25 0 25 50 75 TEMPERATURE (C) 100 125
0
1
4 3 2 INPUT VOLTAGE (V)
5
6
3565 G15
3565 G14
Switch Leakage vs Temperature
600 500 SWITCH LEAKAGE (nA) 400 300 200 100 0 -50 MAIN SWITCH SYNCHRONOUS SWITCH SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 200mA/DIV
Burst Mode Operation
SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 200mA/DIV 4s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 50mA
3565 G17
Pulse Skipping Mode
4s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 5mA
3565 G18
-25
50 25 75 0 TEMPERATURE (C)
100
125
3565 G16
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LTC3565 TYPICAL PERFORMANCE CHARACTERISTICS
otherwise noted. Forced Continuous Mode
SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 200mA/DIV 2s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 80mA
3565 G19
TA = 25C, VIN = 3.6V, fO = 1MHz, unless
Start-Up from Shutdown
RUN 2V/DIV RUN 2V/DIV
Start-Up from Shutdown
VOUT 1V/DIV IL 500mA/DIV 400s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A Burst Mode OPERATION
3565 G20
VOUT 1V/DIV IL 1A/DIV 400s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 1.25A Burst Mode OPERATION
3565 G21
Start-Up from Shutdown with a Prebiased Output (Forced Continuous Mode)
VOUT 1V/DIV VOUT 100mV/DIV AC COUPLED
Load Step
VOUT 100mV/DIV AC COUPLED
Load Step
IL 500mA/DIV
IL 1A/DIV ILOAD 1A/DIV 200s/DIV VIN = 3.6V PREBIASED VOUT = 3V, VOUT = 1.8V ILOAD = 0A
3565 G22
IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A to 1.25A Burst Mode OPERATION
3565 G23
40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 50mA to 1.25A Burst Mode OPERATION
3565 G24
Load Step
VOUT 100mV/DIV AC COUPLED IL 1A/DIV IL 2A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 250mA to 1.25A Burst Mode OPERATION
3565 G25
VOUT Short to Ground
VOUT 1V/DIV
VOUT Short to VIN (Forced Continuous Mode)
VOUT 1V/DIV
IL 500mA/DIV
40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A
3565 G26
40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A
3565 G27
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LTC3565 PIN FUNCTIONS
RT (Pin 1): Timing Resistor Pin. The oscillator frequency is programmed by connecting a resistor from this pin to ground. RUN (Pin 2): Converter Enable Pin. Forcing this pin above 1.5V enables this part, while forcing it below 0.3V causes the device to shut down. In shutdown, the device draws <1A supply current. This pin must be driven; do not float. SYNC/MODE (Pin 3): Combination Mode Selection and Oscillator Synchronization Pin. This pin controls the operation of the device. When tied to SVIN or GND, Burst Mode operation or pulse skipping mode is selected, respectively. If this pin is held at half of SVIN, the forced continuous mode is selected. The oscillation frequency can be synchronized to an external oscillator applied to this pin. When synchronized to an external clock, pulse skip mode is selected. SW (Pin 4): The Switch Node Connection to the Inductor. This pin swings from PVIN to GND. GND (Pin 5): Main Power Ground Pin. Connect to the (-) terminal of COUT, and (-) terminal of CIN.
PIN 1 2 3 4 5 6 7 8 9 10 NAME RT RUN SW GND PVIN SVIN PGOOD VFB ITH DESCRIPTION Timing Resistor Enable Pin Switch Node Main Power Ground Main Power Supply Signal Power Supply Power Good Pin Output Feedback Pin Error Amplifier Compensation -0.3 2.5 0 0 0 0.8 MIN -0.3 -0.3 0 0 0 5.5 5.5 SVIN 1.0 1.5 -0.3 -0.3 -0.3 -0.3 -0.3 6 6 SVIN + 0.3 SVIN + 0.3 SVIN + 0.3 NOMINAL (V) TYP 0.4 ABSOLUTE MAX (V) MAX SVIN SVIN SVIN PVIN MIN -0.3 -0.3 -0.3 -0.3 MAX SVIN + 0.3 SVIN SVIN + 0.3 PVIN + 0.3
PVIN (Pin 6): Main Supply Pin. Must be closely decoupled to GND. SVIN (Pin 7): The Signal Power Pin. All active circuitry is powered from this pin. Must be closely decoupled to GND. SVIN must be greater than or equal to PVIN. PGOOD (Pin 8): The Power Good Pin. This common drain logic output is pulled to GND when the output voltage is not within 7% of regulation. VFB (Pin 9): Receives the feedback voltage from the external resistive divider across the output. Nominal voltage for this pin is 0.6V. ITH (Pin 10): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.4V to 1.4V. Exposed Pad (Pin 11): Ground. Must be soldered to electrical ground on PCB.
SYNC/MODE Mode Select/Synchronization Pin
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LTC3565 BLOCK DIAGRAM
SVIN 7 GND 5 ITH 10 PVIN 6
0.6V
VOLTAGE REFERENCE
ITH LIMIT BCLAMP
PMOS CURRENT COMPARATOR
+ -
+ -
VFB 9
-
ERROR AMPLIFIER VB
+
BURST COMPARATOR SLOPE COMPENSATION 4 SW
0.645V
+ -
OSCILLATOR
+
0.555V PGOOD 8 NMOS COMPARATOR
-
LOGIC
+ - - +
5 GND
REVERSE COMPARATOR 2 RUN 1 RT 3 SYNC/MODE
3565 BD
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LTC3565 OPERATION
The LTC3565 uses a constant frequency, current mode architecture. The operating frequency is determined by the value of the RT resistor or can be synchronized to an external oscillator. To suit a variety of applications, the selectable MODE pin allows the user to trade-off noise for efficiency. The output voltage is set by an external divider returned to the VFB pin. An error amplifier compares the divided output voltage with the reference voltage of 0.6V and adjusts the peak inductor current accordingly. Overvoltage and undervoltage comparators will pull the PGOOD output low if the output voltage is not within 7% of its regulated value. A tripping delay of 40s and untripping delay of 105s ensures PGOOD will not glitch due to transient spikes on VOUT. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle. Current flows through this switch into the inductor and the load, increasing until the peak inductor current reaches the limit set by the voltage on the ITH pin. Then, the top switch is turned off, the bottom switch is turned on, and the energy stored in the inductor forces the current to flow through the bottom switch and the inductor out into the load until the next clock cycle. The peak inductor current is controlled by the voltage on the ITH pin, which is the output of the error amplifier. The output is developed by the error amplifier comparing the feedback voltage, VFB, to the 0.6V reference voltage. When the load current increases, the output voltage and VFB decrease slightly. This decrease in VFB causes the error amplifier to increase the ITH voltage until the average inductor current matches the new load current. The main control loop is shut down by grounding the RUN pin, resetting the internal soft-start. Re-enabling the main control loop by pulling RUN high activates the internal soft-start, which slowly ramps the output voltage over approximately 0.9ms until it reaches regulation. Low Current Operation Three modes are available to control the operation of the LTC3565 at low currents. All three modes automatically switch from continuous operation to the selected mode when the load current is low. To optimize efficiency, the Burst Mode operation can be selected. When the load is relatively light, the LTC3565 automatically switches into Burst Mode operation in which the PMOS switch operates intermittently based on load demand. By running cycles periodically, the switching losses which are dominated by the gate charge losses of the power MOSFETs are minimized. The main control loop is interrupted when the output voltage reaches the desired regulated value. The burst comparator trips when ITH is below approximately 0.5V, shutting off the switch and reducing the power. The output capacitor and the inductor supply the power to the load until ITH rises above approximately 0.5V, turning on the switch and the main control loop which starts another cycle. For lower output voltage ripple at low currents, pulse skipping mode can be used. In this mode, the LTC3565 continues to switch at a constant frequency down to very low currents, where it will eventually begin skipping pulses. Finally, in forced continuous mode, the inductor current is constantly cycled which creates a fixed output voltage ripple at all output current levels. This feature is desirable in telecommunications since the noise is at a constant frequency and is thus easy to filter out. Another advantage of this mode is that the regulator is capable of both sourcing current into a load and sinking current from the output. Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases to 100% which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. Low Supply Operation The LTC3565 incorporates an undervoltage lockout circuit which shuts down the part when the input voltage drops below about 1.9V to prevent unstable operation.
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LTC3565 APPLICATIONS INFORMATION
A general LTC3565 application circuit is shown in Figure 4. External component selection is driven by the load requirement, and begins with the selection of the inductor L1. Once L1 is chosen, CIN and COUT can be selected. Operating Frequency Selection of the operating frequency is a trade-off between efficiency and component size. High frequency operation allows the use of smaller inductor and capacitor values. Operation at lower frequencies improves efficiency by reducing internal gate charge losses but requires larger inductance values and/or capacitance to maintain low output ripple voltage. The operating frequency, fO, of the LTC3565 is determined by an external resistor that is connected between the RT pin and ground. The value of the resistor sets the ramp current that is used to charge and discharge an internal timing capacitor within the oscillator and can be calculated by using the following equation: RT = 1.21 x 106 (fO)-1.2674 (k) where RT is in k and fO is in kHz or can be selected using Figure 1. The maximum usable operating frequency is limited by the minimum on-time and the duty cycle. This can be calculated as: V fO(MAX) 6.67 * OUT (MHz) VIN(MAX) The minimum frequency is limited by leakage and noise coupling due to the large resistance of RT. Inductor Selection The operating frequency, fO, has a direct effect on the inductor value, which in turn influences the inductor ripple current, IL: IL = VOUT V * 1 OUT fO * L V IN
FREQUENCY (kHz)
inductances, but results in higher output ripple voltage, greater core loss and lower output capability. A reasonable starting point for setting ripple current is IL = 0.4 * IOUT(MAX), where IOUT(MAX) is 1.25A. The largest ripple current IL occurs at the maximum input voltage. To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation: L= VOUT VOUT *1 fO * IL V IN(MAX)
The inductor value will also have an effect on Burst Mode operation. The transition from low current operation begins when the peak inductor current falls below a level set by the burst clamp. Lower inductor values result in higher ripple current which causes this to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values will cause the burst frequency to increase.
5000 4500 4000 3500 3000 2500 2000 1500 1000 500 0 0 100 200 300 400 RT (k) 500 600
3565 F01
TA = 25C
Figure 1. Frequency vs RT
Inductor Core Selection Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3565 requires to operate. Table 1
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The inductor ripple current decreases with larger inductance or frequency, and increases with higher VIN or VOUT. Accepting larger values of IL allows the use of lower
10
LTC3565 APPLICATIONS INFORMATION
shows some typical surface mount inductors that work well in LTC3565 applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER Toko A914BYW-1R2M=P3: D52LC A960AW-1R2M=P3: D518LC MAX DC VALUE CURRENT 1.2H 1.2H 2.15A 1.8A 2.1A 2.1A 2.1A 2.05A 3.3A 2.1A 2.3A 2.4A 2.1A 2.2A 2.2A 2.1A 3.0A 3.8A 2.56A 2.80A 2.04A 2.16A 2.6A 3.2A 2.9A 3.2A 1.4A DCR 44m HEIGHT 2mm
diode peak current and average power dissipation so as not to exceed the diode ratings. The main problem with Schottky diodes is that their parasitic capacitance reduces the efficiency, usually negating the possible benefits for LTC3565 circuits. Another problem that a Schottky diode can introduce is higher leakage current at high temperatures, which could reduce the low current efficiency. Remember to keep lead lengths short and observe proper grounding (see Board Layout Considerations) to avoid ringing and increased dissipation when using a catch diode. Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT/VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT (VIN - VOUT ) VIN
46m 1.8mm 43m 1.5mm 45m 1.8mm 47m 49m 22m 80m 70m 72m 2mm 2mm 3mm 1mm 3mm 3mm
DB3015C-1068AS-1R0N 1.0H DB3018C-1069AS-1R0N 1.0H DB3020C-1070AS-1R0N 1.0H A914BYW-2R2M-D52LC 2.2H A915AY-2ROM-D53LC Coilcraft LPO1704-122ML D01608C-222 LP01704-222M Sumida CR32-1R0 CR5D11-1R0 CDRH3D14-1R2 CDRH4D18C/LD-1R1 CDRH4D28C/LD-1R0 CDRH4D28C-1R1 CDRH4D28-1R2 CDRH6D12-1R0 CDRH4D282R2 CDC5D232R2 Taiyo Yuden NPO3SB1ROM N06DB2R2M N05DB2R2M Murata FDK LQN6C2R2M04 MIPW3226DORGM 2.0H 1.2H 2.2H 2.2H 1.0H 1.0H 1.2H 1.1H 1.0H 1.1H 1.2H 1.0H 2.2H 2.2H 1.0H 2.2H 2.2H 2.2H 0.9H
120m 1mm 40m 1.2mm 36m 1.5mm 24m 22m 2mm 3mm 17.5m 3mm 23.6m 3mm 37.5m 1.5mm 23m 3mm 30m 2.5mm 27m 1.8mm 29m 3.2mm 32m 2.8mm 24m 80m 5mm 1mm
where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX ILIM - IL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling, when not using an all ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required ESR to minimize voltage ripple and load step transients. Typically, once the ESR requirement is satisfied, the capacitance
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Catch Diode Selection Although unnecessary in most applications, a small improvement in efficiency can be obtained in a few applications by including the optional diode D1 shown in Figure 2, which conducts when the synchronous switch is off. When using Burst Mode operation or pulse skip mode, the synchronous switch is turned off at a low current and the remaining current will be carried by the optional diode. It is important to adequately specify the
11
LTC3565 APPLICATIONS INFORMATION
is adequate for filtering. The output ripple (VOUT) is determined by: VOUT IL ESR + 1 8fO COUT In most cases, 0.1F to 1F of ceramic capacitors should also be placed close to the LTC3565 in parallel with the main capacitors for high frequency decoupling. Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3565's control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. Refer to Linear Technology Application Note 88 for a detailed discussion of this potential issue. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough to support the load. The time required for the feedback loop to respond is dependent on the compensation components and the output capacitor value. Typically, 3 to 4 cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP, is usually about 2 to 3 times the linear
where f = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. With IL = 0.4 * IOUT(MAX), the output ripple will be less than 100mV at maximum VIN, a minimum COUT of 10F and fO = 1MHz with: ESRCOUT < 150m Once the ESR requirements for COUT have been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement, except for an all ceramic solution. In surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolytic, special polymer, ceramic and dry tantalum capacitors are all available in surface mount packages. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Special polymer capacitors, such as Sanyo POSCAP, offer very low ESR, but have a lower capacitance density than other types. Tantalum capacitors have the highest capacitance density, but it has a larger ESR and it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors have a significantly larger ESR, and is often used in extremely cost-sensitive applications provided that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have the lowest ESR and cost but also have the lowest capacitance density, a high voltage and temperature coefficient and exhibit audible piezoelectric effects. In addition, the high Q of ceramic capacitors along with trace inductance can lead to significant ringing. Other capacitor types include the Panasonic specialty polymer (SP) capacitors.
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12
LTC3565 APPLICATIONS INFORMATION
drop of the first cycle. Thus, a good place to start is with the output capacitor value of approximately: COUT 2.5 IOUT fO * VDROOP will ramp from zero to full scale over a time period of approximately 0.9ms. This prevents the LTC3565 from having to quickly charge the output capacitor and thus supplying an excessive amount of instantaneous current. The LTC3565 can start into a back-biased output in force continuous operation. When the output is pre-biased at either a higher or lower value than the regulated output voltage, the LTC3565 will sink or source current as needed to bring the output back into regulation. However, during soft-start the regulator will always start in pulse skip mode ignoring the mode selected with the SYNC/MODE pin. This prevents the output from discharging to below the regulation point when soft-starting. Mode Selection and Frequency Synchronization The SYNC/MODE pin is a multipurpose pin which provides mode selection and frequency synchronization. Connecting this pin to VIN enables Burst Mode operation, which provides the best low current efficiency at the cost of a higher output voltage ripple. When this pin is connected to ground, pulse skipping operation is selected which provides the lowest output voltage and current ripple at the cost of low current efficiency. Applying a voltage that is half the value of the input voltage results in forced continuous mode, which creates a fixed output ripple and is capable of sinking up to 0.4A. Since the switching noise is constant in this mode, it is also the easiest to filter out. The LTC3565 can also be synchronized to an external clock signal by the SYNC/MODE pin. The internal oscillator frequency should be set to 20% of the external clock frequency to ensure adequate slope compensation, since slope compensation is derived from the internal oscillator. During synchronization, the mode is set to pulse skipping and the top switch turn on is synchronized to the falling edge of the external clock.
More capacitance may be required depending on the duty cycle and load step requirements. In most applications, the input capacitor is merely required to supply high frequency bypassing, since the impedance to the supply is very low. A 10F ceramic capacitor is usually enough for these conditions. Setting the Output Voltage The LTC3565 develops a 0.6V reference voltage between the feedback pin, VFB, and the signal ground as shown in Figure 4. The output voltage is set by a resistive divider according to the following formula: VOUT 0.6V 1+ R2 R1
Keeping the current small (<5A) in these resistors maximizes efficiency, but making them too small may allow stray capacitance to cause noise problems and reduce the phase margin of the error amp loop. To improve the frequency response, a feed-forward capacitor CF may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Shutdown and Soft-Start Pulling the RUN pin high allows an internal soft-start circuit to slowly ramp the output voltage up until regulation. Soft-start prevents surge currents from VIN by gradually ramping the output voltage up during start-up. The output
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13
LTC3565 APPLICATIONS INFORMATION
Checking Transient Response The OPTI-LOOP(R) compensation allows the transient response to be optimized for a wide range of loads and output capacitors. The availability of the ITH pin not only allows optimization of the control loop behavior but also provides a DC coupled and AC filtered closed loop response test point. The DC step, rise time and settling time at this test point truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at the pin. The ITH external components shown in the circuit on page 1 of this data sheet will provide an adequate starting point for most applications. The series R-C filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1s to 10s will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT imVIN C6
mediately shifts by an amount equal to ILOAD * ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine phase margin. The gain of the loop increases with R and the bandwidth of the loop increases with decreasing C. If R is increased by the same factor that C is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. In addition, a feedforward capacitor CF can be added to improve the high frequency response, as shown in Figure 2. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Linear Technology Application Note 76.
OPTI-LOOP is a registered trademark of Linear Technology Corporation.
+
CIN C8
R6 SVIN PVIN LTC3565 RUN SYNC/MODE ITH VFB PGOOD SW
R5 L1 D1 OPTIONAL CF R2 RT GND
3565 F04
PGOOD
+
VOUT COUT C5
CITH
RC CC
R1 RT
Figure 2. LTC3565 General Schematic
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14
LTC3565 APPLICATIONS INFORMATION
Although a buck regulator is capable of providing the full output current in dropout, it should be noted that as the input voltage VIN drops toward VOUT, the load step capability does decrease due to the decreasing voltage across the inductor. Applications that require large load step capability near dropout should use a different topology such as SEPIC, Zeta or single inductor, positive buck/boost. In some applications, a more severe transient can be caused by switching in loads with large (>1F) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power.
1 VIN = 3.6V fO = 1MHz 0.1 POWER LOSS (W)
Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in LTC3565 circuits: 1) LTC3565 VIN current, 2) switching losses, 3) I2R losses, 4) other losses. 1) The VIN current is the DC supply current given in the electrical characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 2) The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3) I2R losses are calculated from the DC resistances of the internal switches, RSW, and external inductor, RL. In continuous mode, the average output current flowing through inductor L is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2(RSW + RL) 4) Other "hidden" losses such as copper trace and internal battery resistances can account for additional efficiency degradations in portable systems. It is very important to include these "system" level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other
Hot Swap is a trademark of Linear Technology Corporation.
0.01
0.001 VOUT = 1.2V VOUT = 1.5V V = 1.2V - 1.8V VOUT OUT = 1.8V 1 10 100 1000 LOAD CURRENT (mA) 10000
3565 F05
0.0001 0.1
Figure 3. Power Loss vs Load Currrent
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15
LTC3565 APPLICATIONS INFORMATION
losses including diode conduction losses during dead-time and inductor core losses, which generally account for less than 2% total additional loss. Thermal Considerations In a majority of applications, the LTC3565 does not dissipate much heat due to its high efficiency. However, in applications where the LTC3565 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3565 from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TRISE = PD * JA where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TRISE + TAMBIENT As an example, consider the case when the LTC3565 is in dropout at an input voltage of 3.3V with a load current of 1A. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) resistance of the P-channel switch is 0.160. Therefore, power dissipated by the part is: PD = IOUT2 * RDS(ON) = 160mW The MSE package junction-to-ambient thermal resistance, JA, will be in the range of about 40C/W. Therefore, the junction temperature of the regulator operating in a 70C ambient temperature is approximately: TJ = 0.16 * 40 + 70 = 76.4C Remembering that the above junction temperature is obtained from an RDS(ON) at 25C, we might recalculate the junction temperature based on a higher RDS(ON) since it increases with temperature. However, we can safely assume that the actual junction temperature will not exceed the absolute maximum junction temperature of 125C. Design Example As a design example, consider using the LTC3565 in a portable application with a Li-Ion battery. The battery provides a VIN = 2.5V to 4.2V. The load requires a maximum of 1.25A in active mode and 10mA in standby mode. The output voltage is VOUT = 2.5V. Since the load still needs power in standby, Burst Mode operation is selected for good low load efficiency. First, calculate the timing resistor for 1MHz operation: RT = 1.21 * 106 (103)-1.2674 = 190.8k Use a standard value of 191k. Next, calculate the inductor value for about 40% ripple current at maximum VIN: L= 2.5V 2.5V *1 = 2H 4.2V 1MHz * 500mA
Choosing the closest inductor from a vendor of 2.2H, results in a maximum ripple current of: IL = 2.5V 2.5V *1 = 460mA 4.2V 1MHz * 2.2H
For cost reasons, a ceramic capacitor will be used. COUT selection is then based on load step droop instead of ESR requirements. For a 5% output droop: COUT 2.5 1.25A = 25F 1MHz * (5% * 2.5V)
The closest standard value is 22F. Since the output impedance of a Li-Ion battery is very low, CIN is typically 22F. In noisy environments, decoupling SVIN from PVIN with an R6/C8 filter of 1/0.1F may help, but is typically not needed.
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16
LTC3565 APPLICATIONS INFORMATION
The output voltage can now be programmed by choosing the values of R1 and R2. To maintain high efficiency, the current in these resistors should be kept small. Choosing 2A with the 0.6V feedback voltage makes R1~300k. A close standard 1% resistor value is 294k then R2 is 931k. The compensation should be optimized for these components by examining the load step response but a good place to start for the LTC3565 is with a 12.1k and 680pF filter. The output capacitor may need to be increased depending on the actual undershoot during a load step. The PGOOD pin is a common drain output and requires a pullup resistor. A 100k resistor is used for adequate speed. The circuit on page 1 of this data sheet shows the complete schematic for this design example. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3565. These items are also illustrated graphically in the layout diagram of Figure 4. Check the following in your layout: 1. Does the capacitor CIN connect to the power VIN (Pin 6) and power GND (Pin 5) as close as possible? This capacitor provides the AC current to the internal power MOSFETs and their drivers. 2. Are the COUT and L1 closely connected? The (-) plate of COUT returns current to PGND and the (-) plate of CIN. 3. The resistor divider, R1 and R2, must be connected between the (+) plate of COUT and a ground line. The feedback signal VFB should be routed away from noisy components and traces, such as the SW line (Pin 4), and its trace should be minimized. 4. Keep sensitive components away from the SW pin. The input capacitor CIN, the compensation capacitor CC and CITH and all the resistors R1, R2, RT, and RC should be routed away from the SW trace and the inductor L1. The SW pin pad should be kept as small as possible. 5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the GND pin at one point. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to one of the input supply rails: PVIN, SVIN or GND.
CC RT 1 2 RC 10 9 8 7 6
CITH R1 R2 C4
RT RUN LTC3565
ITH VFB PGOOD SVIN PVIN
L1 VOUT COUT
3 SYNC/MODE 4 SW 5 GND
R5 VIN
3565 F06
CIN BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 4. LTC3565 Layout Diagram (See Board Layout Checklist)
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17
LTC3565 TYPICAL APPLICATION
General Purpose Buck Regulator Using Ceramic Capacitors
VIN 2.5V TO 5.5V C1 22F PVIN LTC3565 SYNC/MODE PS RS2 1M R3 12.1k C3 680pF ITH GND R4 191k R1A 147k R1B 196k R5 100k PGOOD
SVIN BM FC RS1 1M
PGOOD SW
L1 2.2H R2 294k
VFB SHDN/RT 1.8V 1.5V 1.2V C4 22pF R1C 294k
VOUT 1.8V/1.5V/1.2V AT 1.25A
C2 22F
3565 TA02a
NOTE: IN DROPOUT, THE OUTPUT TRACKS THE INPUT VOLTAGE C1, C2: TAIYO YUDEN JMK325BJ226MM L1: TOKO A914BYW-2R2M (D52LC SERIES)
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 1 VIN = 3.6V VOUT = 1.2V fO = 1MHz 10 100 1000 OUTPUT CURRENT (mA) 10000
3565 TA02b
Burst Mode OPERATION VOUT 100mV/DIV AC COUPLED PULSE SKIP IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.2V ILOAD = 100mA TO 1.25A Burst Mode OPERATION
3565 TA02c
FORCED CONTINUOUS
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 100mA TO 1.25A PULSE SKIPPING MODE
3565 TA02d
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18
LTC3565 PACKAGE DESCRIPTION
DD Package 10-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1699)
R = 0.115 TYP 6 0.675 0.05 10 0.38 0.10
3.50 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS PIN 1 TOP MARK (SEE NOTE 6)
3.00 0.10 (4 SIDES)
1.65 0.10 (2 SIDES)
(DD) DFN 1103
5 0.200 REF 0.75 0.05 2.38 0.10 (2 SIDES)
1 0.25 0.05 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2). CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
MSE Package 10-Lead Plastic MSOP Exposed Die Pad ,
(Reference LTC DWG # 05-08-1664 Rev B)
2.794 0.102 (.110 .004)
0.889 0.127 (.035 .005)
3.00 0.102 (.118 .004) (NOTE 3)
BOTTOM VIEW OF EXPOSED PAD OPTION
10 9 8 7 6 0.497 0.076 (.0196 .003) REF
1
2.06 0.102 (.081 .004)
1.83 0.102 (.072 .004)
5.23 (.206) MIN
2.083 0.102 3.20 - 3.45 (.082 .004) (.126 - .136)
0.254 (.010) DETAIL "A" 0 - 6 TYP
4.90 0.152 (.193 .006)
3.00 0.102 (.118 .004) (NOTE 4)
0.50 0.305 0.038 (.0197) (.0120 .0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT
GAUGE PLANE 0.53 0.152 (.021 .006) DETAIL "A" 0.18 (.007) SEATING PLANE 1.10 (.043) MAX
12345 0.86 (.034) REF
10
0.17 -0.27 (.007 - .011) TYP
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.50 (.0197) BSC
0.1016 0.0508 (.004 .002)
MSOP (MSE) 0307 REV B
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3565 TYPICAL APPLICATION
VIN 2.5V TO 4.2V
1mm Height, 2MHz, Li-Ion to 1.8V Converter
C1 10F R5 100k PVIN SVIN LTC3565 RUN SYNC/MODE ITH R3 13.3k C3 470pF GND VFB RT R4 80.6k R1 232k R2 464k C1, C2: TAIYO YUDEN JMK107BJ106MA L1: FDK MIPW3226DORGM PGOOD SW L1 0.9H PGOOD C4 22pF C2 10F x2 VOUT 1.8V AT 1.25A
3565 TA04a
Efficiency vs Output Current
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 VOUT = 1.8V fO = 2MHz 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10000
3565 TA04b
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 50mA TO 1.25A
3565 TA04c
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV
10 100 1000 OUTPUT CURRENT (mA)
40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 250mA TO 1.25A
3565 F04d
RELATED PARTS
COMMENTS 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20A, ISD < 1A, ThinSOTTM 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, LTC3407A/LTC3407AB Dual 600mA/800mA (IOUT ), 1.5MHz/2.25MHz Synchronous Step-Down DC/DC Converters IQ = 40A, ISD < 1A, MS10E, DFN LTC3410/LTC3410B 300mA (IOUT ), 2.25MHz Synchronous Step-Down DC/DC Converters 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26A, ISD < 1A, SC70 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, LTC3411A 1.25A (IOUT ), 4MHz Synchronous Step-Down DC/DC Converter IQ = 60A, ISD < 1A, MS10, 3mm x 3mm DFN 96% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, LTC3412A 3A (IOUT ), 4MHz Synchronous Step-Down DC/DC Converter IQ = 62A, ISD < 1A, TSSOP16E, 4mm x 4mm QFN LTC3531/LTC3531-3 200mA (IOUT ), 1.5MHz Synchronous Buck-Boost DC/DC Converters 95% Efficiency, VIN: 1.8V to 5.5V, VOUT(MIN) = 2V to 5V, IQ = 16A, ISD < 1A, ThinSOT, DFN LTC3531-3.3 95% Efficiency, VIN: 2.4V to 5.5V, VOUT(MIN) = 2.4V to LTC3532 500mA (IOUT ), 2MHz Synchronous Buck-Boost DC/DC Converter 5.25V, IQ = 35A, ISD < 1A, MS10, DFN LTC3542 500mA (IOUT ), 2.25MHz Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 26A, ISD < 1A, 2mm x 2mm DFN LTC3544 Quad 300mA + 2x 200mA + 100mA, 2.25MHz Synchronous 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 70A, ISD < 1A, 3mm x 3mm QFN Step-Down DC/DC Converter LTC3547/LTC3547B Dual 300mA, 2.25MHz Synchronous Step-Down DC/DC Converters 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD < 1A, 2mm x 3mm DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, LTC3548/LTC3548-1 Dual 400mA/800mA, (IOUT ), 2.25MHz Synchronous Step-Down DC/DC Converters IQ = 40A, ISD < 1A, MS10E, DFN LTC3548-2 LTC3560 800mA (IOUT ), 2.25MHz Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 16A, ISD < 1A, ThinSOT ThinSOT is a trademark of Linear Technology Corporation.
3565f
PART NUMBER LTC3406/LTC3406B
DESCRIPTION 600mA (IOUT ), 1.5MHz Synchronous Step-Down DC/DC Converters
20 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0109 * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2009

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